a) Field of the Invention
The present invention concerns DC to DC and single phase fed AC to DC switching power converters, and in particular, concerns switching power converters, having an output power of up to 5 KW (Kilo Watts), employing a novel lossless zero-voltage-switching (ZVS) passive snubber network. The proposed passive snubber network improves efficiency, power density, and transient performance, reduces switching losses and electro-magnetic interference (EMI), and permits fixed frequency operation of switching power converters. The proposed passive snubber network also reduces and/or eliminates large peak currents and reverse recovery current spikes which normally occur in known switching power converters. The proposed passive snubber network may be used in various switching power converter topologies such as boost, buck, forward, and flyback power converters.
b) Prior Art
In the past, power conversion, such as DC to DC power conversion for example, was typically performed by hard switched, pulse width modulating (PWM) circuits such as the "boost" power converter shown in FIG. 1, for example. This known power converter includes a controllable switch S.sub.1 (such as a transistor (e.g., a MOSFET), for example) which can be provided with a fixed frequency switching signal from a controller. By varying the duty cycle of the switching signal, the output current of the power converter is controlled. This is known as pulse width modulation (or PWM) control.
Specifically, the conventional boost power converter of FIG. 1 includes a series connection of an inductor L.sub.1, a diode D.sub.1, and a capacitor C.sub.1 coupled with the input voltage V.sub.IN. The anode of the diode D.sub.1 is coupled with the inductor L.sub.1 while the cathode of the diode D.sub.1 is coupled with the capacitor C.sub.1. A load R.sub.L to be supplied with an output voltage is coupled across the capacitor C.sub.1. A controllable switch S.sub.1, such as a MOSFET for example, includes a first terminal coupled with a node between the inductor L.sub.1 and the anode of the diode D.sub.1 and a second terminal coupled with a lower potential terminal of the input voltage source V.sub.IN.
In the conventional boost converter of FIG. 1, when the controllable switch S.sub.1 is open (i.e., blocking), current flows through the inductor L.sub.1 and the diode D.sub.1. Since the uncharged capacitor C.sub.1 initially appears as a short circuit (because it will draw current), the current flowing through the inductor L.sub.1 and the diode D.sub.1 will charge the capacitor C.sub.1. When the controllable switch S.sub.1 is subsequently closed (conducting), current from the input voltage V.sub.IN will flow through the inductor L.sub.1 and the controllable switch S.sub.1 to ground (or to the negative terminal of the input voltage supply). Assuming that there are no losses in the inductor L.sub.1, equating the volt-seconds across the inductor L.sub.1 to zero, and ignoring the turn-on voltage of the diode D.sub.1, the output voltage V.sub.OUT can be determined from the following relationship: ##EQU1## where D is the duty cycle of the switching signal. Thus, assuming that a fixed frequency switching signal is provided to the controllable switch S.sub.1, the higher the duty cycle of the switching signal, the higher the output voltage supplied across the output load R.sub.L.
The simple design of the known boost converter of FIG. 1 is based on an assumption that existing power switches closely approximate ideal switches, i.e., that the transitions from opened (i.e., blocking) to closed (i.e., conducting) and closed to opened occur instantaneously. Unfortunately, this assumption is not particularly accurate. Indeed, this assumption, and the power converter topologies it has spawned, are responsible for serious limitations in the performance of the conventional switching converters because the non-ideal (i.e., non-instantaneous) switching characteristic causes switching power (P=I.sub.switch *V.sub.switch) losses.
Specifically, one of the most important characteristics of a DC to DC power converter is that it has a high power density. To possess a high power density, the controllable switch of the power converter must operate at relatively high frequencies. Thus, conventional pulse width modulation power converters, such as the conventional boost power converter discussed above, are disadvantageous because increasing the switching frequencies to achieve higher power densities will cause an increase in switching losses.
In conventional hard switching PWM converters operating at frequencies below 100 KHz and at power levels up to 5 KW, diode stored charge, diode reverse recovery, and device switching losses are reportedly the biggest problems.
Diode reverse recovery in the conventional hard switching boost power factor converter presents a significant limitation because it generates substantial EMI and limits the power conversion frequency and efficiency, particularly in the 3 to 5 KW power range. Specifically, in the conventional hard switching boost converter of FIG. 1, when the switch S.sub.1 is closed, the current through the switch S.sub.1 increases to the level of the current through the inductor L.sub.1. At this point, the current through the diode D.sub.1 decreases until the diode D.sub.1 no longer conducts. At this time, any charge stored on the diode D.sub.1 is removed via switch S.sub.1. As the charge is being removed from the diode D.sub.1, the current through the switch S.sub.1 continues to rise, often to a value of more than twice the inductor current level. The combination of high peak current, high dI/dt (current rate of change), and high dV/dt (voltage rate of change) when the voltage of the switch S.sub.1 approaches the level of the lower potential terminal of V.sub.in (or ground), creates significant unwanted RFI/EMI noise and considerably stresses the switch S.sub.1.
The problems of conventional hard switching PWM power converters (such as boost, buck, forward, and flyback power converters) are explained in greater detail in the article D. M. Divan, "Soft Switching Converters: A Review," Soft Switching Converters: Topologies, Design, and Control: Summary of Publications 1986-1990: Wisconsin Electric Machines and Power Electronics Consortium, pp. 1-51 (1990) (hereinafter referred to as "the Divan article" and incorporated herein by reference).
In response to the problems associated with hard switching PWM power converters described above, designers have proposed the use of reactive snubber networks to divert energy that would be dissipated during switching transitions by "trapping" that energy, thereby permitting "soft-switching" of the controllable switch (i.e., switching when little or no voltage appears across the switch and/or when little or no current is flowing through the switch thereby reducing switching stresses). There are two types of such soft-switching converters: (1) zero current switching (ZCS) converters in which opening (switch-off) and closing (switch-on) of the controllable switch occur with no current in the controllable switch; and (ii) zero voltage switching (ZVS) converters in which opening (switch-off) and closing (switch-on) of the controllable switch occur with no voltage across the switch.
Zero current switching (ZCS) is accomplished generally by employing a purely inductive snubber. Zero voltage switching (ZVS) on the other hand is accomplished by employing a purely capacitive snubber having an anti-parallel diode. With ZVS snubber circuits, closing (switch-on) occurs only when the anti-parallel diode is conducting; opening (switch-off) losses decrease with increasing capacitance. Unfortunately, to cause conduction in the anti-parallel diode before turning on the switch, additional circuitry is required to discharge the capacitor or an external resistor is required to dissipate the energy stored in the capacitor during a turn-on part of the switching cycle. Thus, although the controllable switch can operate at elevated frequencies because switching stresses are reduced, the power losses are merely shifted from the controllable switch to the dissipating resistor. Furthermore, if a snubber circuit is not properly designed, it will present a low impedance to the switch when it is turned on and off (closed and opened) which results in a large current spikes.
To solve the power dissipation problem, designers have developed "lossless" soft switching power converters in which the snubber networks are reset by means of inherent circuit operation. These lossless soft switching power converters "recirculate" the energy stored by the reactive snubbers to accomplish lossless operation. Other "lossless" soft switching power converters have snubber networks which are reset by using additional auxiliary switches in conjunction with reactive elements. Unfortunately, this way of eliminating spikes requires extra power handling components. These additional power handing components add size, weight, and cost to the power conversion system. Moreover, they often severely reduce overall system efficiency since the RMS input current is high.
An example of a known zero voltage switching (ZVS) boost power converter (See e.g., J. Bazinet et al., "Analysis and Design of a Zero Voltage Transition Power Factor Correction Circuit," IEEE Applied Power Electronics Conference (APEC), pp. 591-597 (1994)) is illustrated in FIG. 2. As shown in FIG. 2, this known ZVS boost power converter is a modification of the hard switching boost converter of FIG. 1. Specifically, a capacitor C.sub.2 is arranged across the switch S.sub.1. A first series circuit, including an inductor L.sub.2, a diode D.sub.2 and a second switch S.sub.2, is also arranged in parallel with the switch S.sub.1. Further, a second series circuit, including a diode D.sub.3, a diode D.sub.4, and a resistor R.sub.1 is arranged in parallel with the capacitor C.sub.1. The first and second series circuits are electrically coupled, from a node between the inductor L.sub.2 and the diode D.sub.2 of the first series circuit to a node between the diodes D.sub.3 and D.sub.4 of the second series circuit.
The operation of the known ZVS boost power converter is explained below. To achieve zero voltage switching of the switch S.sub.1, the auxiliary switch S.sub.2 is turned-on (i.e., closed) near the end of the time switch S.sub.1 is off (i.e., not conducting). Then, the current in the inductor L.sub.2 increases until it reaches the level of the current in the input inductor L.sub.1. Simultaneously, the capacitor C.sub.2 and the inductor L.sub.2 create a resonance thereby reducing the voltage across the switch S.sub.1 to zero before the switch S.sub.1 is turned-on (closed). The diode D.sub.1 is turned off (opened) without the problem of a high reverse recovery current passing through the switch S.sub.1. The capacitor C.sub.2 minimizes the voltage across the switch S.sub.1 to a very low value during turn-off (opening).
Unfortunately, this known ZVS boost power converter requires an active circuit element; namely, the second switch S.sub.2. Being an active element, the second switch S.sub.2 requires additional supporting circuitry (e.g., a base drive circuit) and introduces additional losses. The turn-off (opening) losses of the switch S.sub.2 are significant because the inductor L.sub.2 and the switch S.sub.2 are carrying the load current before the switch S.sub.2 is turned off (opened). Similarly, during the turn-on (closing) of the switch S.sub.2, the capacitor C.sub.2 will discharge through L.sub.2 and S.sub.2 thereby causing additional power dissipation. Therefore: (i) the energy stored in the parasitic capacitance of the switch S.sub.2 dissipates in the switch S.sub.2 during the turn-on (closing); (ii) the switch S.sub.2 experiences substantial turn-off (opening) losses because before the turn-off (opening) of the switch S.sub.2, it carries the full current of the inductor L.sub.1 ; (iii) the inductor L.sub.2 must be designed to limit any reverse recovery current spike from the diode D.sub.1 during the turn-on (closing) of the switch S.sub.2 ; (iv) the tailing effect of IGBTs (Insulated Gate Bipolar Transistors) during turn-off causes difficulties when using IGBTs in power converters having a power range of 3 to 5 KW.
Thus, the above mentioned known snubber circuits either: (a) use an active auxiliary switch in conjunction with the reactive elements (see e.g., the known ZVS boost power converter of FIG. 2) to relieve the voltage and current switching stresses of the controllable switch; or (b) use the controllable switch to provide energy to the circuit, that is, the energy used by the snubber circuit is drawn from the input and returned to the input.
Other known lossless power converters include resonant switching converters. These power converters incorporate reactive elements (capacitors and inductors) in conjunction with the switching device. The output voltage of these circuits is controlled by varying the operating frequency of the controllable switches. These circuits advantageously have low semiconductor switching losses and operate with sinusoidal waveforms. Unfortunately, resonant power converters exhibit increased component count, increased switching currents (peak and RMS) and require wide operating frequency variations to maintain a constant output voltage. Thus, resonant switching converters are relatively expensive, require relatively complex switching control circuitry, and eliminate switching losses at the expense of conducting losses.
A further example of a zero voltage switching (ZVS) quasi-resonant boost power converter is illustrated on page 14 of the Divan article. In this zero voltage switching (ZVS) boost power converter, a capacitor, along with an anti-parallel diode, are coupled across the controllable switch S.sub.1 and an inductor is placed in series with the controllable switch S.sub.1. (See phantom lines in FIG. 1.) The operation of this power converter is explained below. During the turn-off of the switch S.sub.1, the inductor L.sub.2 and the capacitor C.sub.2 cause a resonance, thereby making the voltage across the switch S.sub.1 nearly zero. Moreover, the switch S.sub.1 is closed when its anti-parallel diode D.sub.2 is conducting. Thus, the current is zero during turn-on (closing) of the switch S.sub.1. However, the current flowing through the switch S.sub.1 is sinusoidal. Thus, the peak and RMS (root mean square) currents are increased. Consequently, as stated above, the quasi-resonant power converter eliminates switching losses at the expense of conduction losses. Furthermore, a wide range of operating switching frequencies is required to maintain a constant output voltage.
Therefore, although these known lossless boost power converters have better characteristics than the hard switching converters and non-lossless soft-switching boost power converters, certain disadvantages remain. Specifically, the known ZVS boost power converter requires an additional auxiliary switch which requires additional support circuitry (e.g., a base drive circuit) and which introduces additional parasitic losses. Furthermore, the turn-off (opening) of the additional switch is not lossless as explained earlier. On the other hand, although resonant switching power converters eliminate switching losses, they introduce conducting losses. Moreover, as stated above, the output voltage in the resonant switching converters is controlled by frequency modulation. Therefore, this circuit requires additional circuitry to carry out such frequency modulation. Accordingly, an improved switching power converter employing an improved snubber network is needed.
The improved snubber network used by such a switching power converter should be a passive network; that is, the snubber network should include only resistors (not required in the present invention), capacitors, inductors, and diodes. The improved snubber network should use relatively inexpensive components. Moreover, the improved snubber network should enable the switching power converter to operate at higher switching frequencies to increase power density and to permit smaller components to be used. The improved snubber network should also eliminate reverse recovery current spikes in the switching power converter. The improved lossless soft switching power converter using the improved snubber network should also limit peak device voltage and current stresses, limit peak capacitor voltages, limit RMS currents in all components, have low sensitivity to second order effects (particularly when high switching frequencies are used), not require complex control circuitry, be fault tolerant, and have low electro-magnetic interference (EMI) and low radio frequency interference (FI). Lastly, the improved snubber network should be adaptable for use in various switching power converter topologies such as boost, buck, forward, and flyback power converters.